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| (R) SP6121 Ideal for 2A to 10A, Small Footprint, DC-DC Power Converters Low Voltage, Synchronous Step Down PWM Controller FEATURES Optimized for Single Input Voltage - 3V to 5.5V 8 PDRV VCC 1 High Efficiency: Greater than 95% possible SP6121 GND 2 7 NDRV Accurate, 500kHz Fixed Frequency Operation 6 ISET VFB 3 8 Pin SOIC Fast Transient Response 4 5 ISENSE COMP 500A, IQ (25A in Shutdown) Internal, 0.4 V/ms, Soft Start Circuit Precision 1% Reference Now Available in Lead Free Packaging Resistor Programmable Output Voltage APPLICATIONS Lossless Adjustable Current Limit with Supply Bias for High Side RDS(ON) Sensing - DSP 0% to 100% Duty Cycle Range - Microprocessor Core High Side PMOS Switch Negates Need for - I/O & Logic External Charge Pump Video Cards Output Over Voltage Protection Board Level Supply in Hiccup Mode Current Limit Protection Distributed Power Systems DESCRIPTION The SP6121 is a fixed frequency, voltage mode, synchronous PWM controller designed to work from a single 5V or 3.3V input supply, providing excellent AC and DC regulation for high efficiency power conversion. The operating frequency is internally set at 500kHz, permitting the use of small, surface mount inductors and capacitors. Requiring only few external components, the SP6121 packaged in an 8-pin SOIC, is especially suited for low voltage applications where cost, small size and high efficiency are critical. With its low voltage capability and inherent 100% duty cycle operation, the SP6121 allows low dropout operation in the event of a low input supply voltage condition. TYPICAL APPLICATION CIRCUIT RVCC 5 (R) RSET 2.3k Q1 1.8H L1 Q2 DS CIN 47F Ceramic 6.3V 3.3V VIN CB 3.3F VCC GND VFB COMP RZ 10k CZ 3.7nF PDRV SP6121 U1 NDRV ISET ISENSE 1.9V 8A VOUT COUT 470F x 2 R1 5.2k R2 10k CP 50pF Q1 = FAIRCHILD FDS6375 Q2 = FAIRCHILD FDS6690A DS = STMICROELECTRONICS STPS2L25BU Date: 5/25/04 L1 = PANASONIC ETQ-P6F1R6SFA COUT = SANYO 4TPB470M (c) Copyright 2004 Sipex Corporation SP6121 Low Voltage, Synchronous Step Down PWM Controller 1 ABSOLUTE MAXIMUM RATINGS VCC ...................................................................................................... 7V All other pins ............................... -0.3V to VCC+0.3V Peak Output Current < 10s PDRV, NDRV ....................................................... 2A Storage Temperature ...................... -65C to 150C Lead Temperature (Soldering, 10 sec) .......... 300C ESD Rating ............................................... 2kV HBM These are stress ratings only and functional operation of the device at these ratings or any other above those indicated in the operation sections of the specifications below is not implied. Exposure to absolute maximum rating conditions for extended periods of time may affect reliability. ELECTRICAL SPECIFICATIONS Unless otherwise specified: 0C < TA < 70C, 3.0V < VCC < 5.5V, CCOMP = 22nF, CPDRV = CNDRV = 3.3nF, VFB = 1.25V, ISET = ISENSE = VCC, GND=0V PARAMETER QUIESCENT CURRENT VCC Supply Current VCC Supply Current (Disabled) ERROR AMPLIFIER Error Amplifier Transconductance COMP Sink Current COMP Source Current COMP Output Impedance VFB Input Bias Current ERROR AMPLIFIER REFERENCE Initial Accuracy Error Amplifier Reference over Line, Load and Temperature OSCILLATOR & DELAY PATH Internal Oscillator Frequency Maximum Duty Cycle Minimum Duty Cycle Minimum PDRV Pulse Width CURRENT LIMIT Internal Current Limit Threshold ISET Sink Current Current Limit Threshold and ISET Temperature Coefficient Current Limit Time Constant ISENSE Input Bias Current SOFT START, SHUTDOWN, UVLO Internal Soft Start Slew Rate Internal Soft Start Delay Time COMP Discharge Current COMP Clamp Voltage COMP Clamp Current MIN TYP MAX UNITS CONDITIONS - 0.5 25 600 1.0 60 mA A s No Switching COMP = 0V 15 15 35 35 3 100 65 65 A A M nA VFB = 1.35V, COMP=0.8V, No Faults VFB =1.15V, COMP=1.8V 1.238 1.225 1.250 1.250 1.262 1.275 V V Trimmed with Error Amp in Unity Gain 440 100 - 500 100 560 0 kHz % % ns COMP = 2V COMP = 0.8V VCC > 4.5V, Ramp up COMP voltage until PDRV starts switching VISET - VISENSE, TA = 25C VISET =5V, TA = 25C 125 25 160 30 0.33 15 195 35 mV A %/C s - 0.4 1.5 100 nA V/ms ms A Measured at COMP pin on the transition from shutdown COMP charging to PDRV switching COMP = 0.5V, Fault Initiated VFB = 1.3V COMP = 0.5V, VFB =1.15V 150 0.6 300 0.7 100 0.8 V A Date: 5/25/04 SP6121 Low Voltage, Synchronous Step Down PWM Controller (c) Copyright 2004 Sipex Corporation 2 ELECTRICAL SPECIFICATIONS: Continued Unless otherwise specified: 0C < TA < 70C, 3.0V < VCC < 5.5V, CCOMP = 22nF, CPDRV = CNDRV = 3.3nF, VFB = 1.25V, ISET = ISENSE = VCC, GND=0V PARAMETER Shutdown Threshold Voltage Shutdown Input Pull-up Current VCC Start Threshold VCC Stop Threshold VCC Hysteresis GATE DRIVERS PDRV Rise Time PDRV Fall Time NDRV Rise Time PDRV Fall Time PDRV to NDRV Non-Overlap Time NDRV to PDRV Non-Overlap Time MIN 0.2 2.69 2.59 - TYP 0.3 5 2.79 2.69 100 40 40 40 40 80 50 MAX UNITS CONDITIONS 0.4 2.89 2.79 110 110 110 110 V A V V mV ns ns ns ns ns ns VCC > 4.5V VCC > 4.5V VCC > 4.5V VCC > 4.5V VCC > 4.5V VCC > 4.5V Measured at COMP Pin COMP = 0.2V, Measured at COMP pin SOFT START, SHUTDOWN, UVLO: continued PIN DESCRIPTION PIN NO. 1 2 3 PIN NAME DESCRIPTION VCC GND VFB Positive input supply for the control circuitry and gate drivers. Properly bypass this pin to GND with a low ESL/ESR ceramic capacitor. Ground pin. Both power and control circuitry of the IC is referenced to this pin. Feedback Voltage Pin. It is the inverting input of the Error Amplifier and serves as the output voltage feedback point for the buck converter. The output voltage is sensed and can be adjusted through an external resistor divider. Output of the Error Amplifier. It is internally connected to the non-inverting input of the PWM comparator. A lead-lag network is typically connected to the COMP pin to compensate the feedback loop in order to optimize the dynamic performance of the voltage mode control loop. Sleep mode can be invoked by pulling the COMP pin below 0.2V with an external open-drain or open-collector transistor. Supply current is reduced to 25A (typical) in shutdown. An internal 5A pull-up ensures start-up. Current Limit Sense pin. Connect this pin to the switching node at the junction between the two external power MOSFET transistors. This pin monitors the voltage dropped across the RDS(ON) of the high side P-channel MOSFET while it is conducting. When this drop exceeds the sum of the voltage programmed through the ISET pin plus the internal 160mV threshold, the overcurrent comparator sets the fault latch and terminates the output pulses. The controller stops switching and goes through a hiccup sequence. This prevents excessive power dissipation in the external power MOSFETs during an overload condition. An internal delay circuit prevents that very short and mild overload conditions, that could occur during a load transient, activate the current limit circuit. Current Limit Threshold pin. An external resistor connected between this pin and the source of the high side P-channel MOSFET adds to the internal current limit threshold of 160mV. If a current limit threshold in excess of 160mV is required, the external programming resistor can properly be chosen based on the internal 30A pull down current available on the ISET pin. Both this 30A current source and the 160mV built-in current limit threshold have a positive temperature coefficient to provide first order correction for the temperature coefficient of the external P-channel MOSFET's RDS(ON). High current driver output for the low side MOSFET switch. It is always low if PDRV is low or during a fault. High current driver output for the high side MOSFET switch. It is always high if NDRV is high or during a fault. SP6121 Low Voltage, Synchronous Step Down PWM Controller (c) Copyright 2004 Sipex Corporation 4 COMP 5 ISENSE 6 ISET 7 8 NDRV PDRV Date: 5/25/04 3 FUNCTIONAL DIAGRAM 1V + - DRIVER ENABLE Reference 1.25V SHUTDOWN FAULT 0.4V/ms SOFTSTART GM ERROR AMP 300mV 5A VCC + RESET Dominant S PWM Logic 8 PDRV Synchronous Driver 7 NDRV + - PWM COMP + VFB 3 COMP 4 VCC 1 UVLO Q R 2 GND 2.79V ON 2.69V OFF ISET 6 PDRV 30A (3300 ppm/C) ISENSE 5 750mV RAMP F = 500kHz + Reset Dominant S Q Over Current (Gated S&H) SHUTDOWN R FAULT COMP + X 3.3 + 530mV (3300 ppm/C) - THEORY OF OPERATION General Overview The SP6121 is a constant frequency, voltage mode, synchronous PWM controller designed for low voltage, DC/DC step down converters. It is intended to provide complete control for a high power, high efficiency, precisely regulated output voltage from a highly integrated 8-pin solution. The internal free-running oscillator accurately sets the PWM frequency at 500kHz without requiring any external elements and allows the use of physically small, low value external components without compromising performance. A transconductance amplifier is used for the error amplifier, which compares an attenuated sample of the output voltage with a precision reference voltage. The output of the error amplifier (COMP), is compared to a 0.75V peak-to-peak ramp waveform to provide PWM control. The COMP pin provides access to the output of the error amplifier and allows the use of external components to stabilize the voltage loop. High efficiency is obtained through the use of synchronous rectification. Synchronous regulators replace the catch diode in the standard buck converter with a low R DS(ON) N-channel Date: 5/25/04 MOSFET switch allowing for significant efficiency improvements. The SP6121 includes two fast MOSFET drivers with internal non-overlap circuitry and drives a complementary pair of power transistors, P-channel on the high side, and N-channel on the low side. The use of a Pchannel high side device minimizes complexity and external component count by eliminating the need for a charge pump that would otherwise be required to fully enhance an N-channel device. It also allows inherent 100% duty cycle for low dropout operation in the event of a low input supply voltage condition. The SP6121 includes an internal 0.4V/ms softstart circuit that provides controlled ramp up of the output voltage, preventing overshoot and inrush current at power up. Current limiting is implemented by monitoring the voltage drop across the RDS(ON) of the high side P-channel MOSFET while it is conducting, thereby eliminating the need for an external sense resistor. The over-current comparator has a built-in threshold of 160mV that can be programmed to higher values using a single external resistor, connected to the ISET pin, whose (c) Copyright 2004 Sipex Corporation SP6121 Low Voltage, Synchronous Step Down PWM Controller 4 THEORY OF OPERATION: continued value is selected to match the MOSFET characteristics. When the over-current threshold is exceeded, the over-current comparator sets the fault latch and terminates the output pulses. The controller stops switching and goes through a hiccup sequence. This prevents excessive power dissipation in the external power MOSFETs during an overload condition. An internal delay circuit prevents that very short and mild overload conditions, that could occur during a load transient, activate the current limit circuit. A low power sleep mode can be invoked in the SP6121 by externally forcing the COMP pin below 0.3V. Quiescent supply current in sleep mode is typically less than 25A. An internal 5A pull-up current at the COMP pin brings the SP6121 out of shutdown mode. The SP6121 also includes under-voltage lockout and over-voltage protection. Output overvoltage protection is achieved by turning off the high side switch, and turning on the low side Nchannel MOSFET full time. Enable Low quiescent mode or "Sleep Mode" is initiated by pulling the COMP pin below 0.3V with an external open-drain or open-collector transistor. Supply current is reduced to 25A (typical) in shutdown. On power-up, assuming that VCC has exceeded the UVLO start threshold (2.79V), an internal 5A pull-up current at the COMP pin brings the SP6121 out of shutdown mode and ensures start-up. During normal operating conditions and in absence of a fault, an internal clamp prevents the COMP pin from swinging below 0.6V. This guarantees that during mild transient conditions, due either to line or load variations, the SP6121 does not enter shutdown unless it is externally activated. During Sleep Mode, the high side and low side MOSFETs are turned off and the internal soft start voltage is held low. UVLO Assuming that there is not shutdown condition present, then the voltage on the VCC pin determines operation of the SP6121. As VCC rises, the UVLO block monitors VCC and keeps the high side and low side MOSFETS off and the internal SS voltage low until VCC reaches 2.79V. Date: 5/25/04 If no faults are present, the SP6121 will initiate a soft start when VCC exceeds 2.79V. Hysteresis (about 100mV) in the UVLO comparator provides noise immunity at start-up. Soft Start Soft start is required on step-down controllers to prevent excess inrush current through the power train during start-up. Typically this is managed by sourcing a controlled current into a timing capacitor and then using the voltage across this capacitor to slowly ramp up either the error amp reference or the error amp output (COMP). The control loop creates narrow width driver pulses while the output voltage is low and allows these pulses to increase to their steady-state duty cycle as the output voltage increases to its regulated value. As a result of controlling the inductor volt*second product during startup, inrush current is also controlled. In the SP6121 the duration of the soft-start is controlled by an internal timing circuit that provides a 0.4V/ms slew-rate, which is used during start-up and over-current to set the hiccup time. The SP6121 implements soft-start by ramping up the error amplifier reference voltage providing a controlled slew-rate of the output voltage, thereby preventing overshoot and inrush current at power up. The presence of the output capacitor creates extra current draw during startup. Simply stated, dVOUT/dt requires an average sustained current in the output capacitor and this current must be considered while calculating peak inrush current and over current thresholds. An approximate expression to determine the excess inrush current due to the dVOUT/dt of the output capacitor COUT is: ICOUT = COUT*(0.4 V/ms) * VOUT 1.25 As Figure 1 shows, the SS voltage controls a variety of signals. First, provided all the external fault conditions are removed, an internal 5A pull-up at the COMP pin brings the SP6121 out of shutdown mode. The internal timing circuit is then activated and controls the rampup of the error amp reference voltage. The COMP pin is pulled to 0.7V by the internal (c) Copyright 2004 Sipex Corporation SP6121 Low Voltage, Synchronous Step Down PWM Controller 5 THEORY OF OPERATION: continued clamp and then gradually charges preventing the error amplifier from forcing the loop to maximum duty cycle. As the COMP voltage crosses about 1V (valley voltage of the PWM ramp), the driver begins to switch the high side MOSFET with narrow pulses in an effort to keep the converter output regulated . The SP6121 operates at low duty cycle as the COMP voltage increases above 1V. As the error amp reference ramps upward, the driver pulses widen until a steady state value is reached and the output voltage is regulated to the final value ending the soft start charge cycle. COMP 1V 0.7 V 0.3 V 0V Internal SS Voltage dVss/dt = 0.4V/ms Error Amp Reference Voltage 1.25V VOUT = V(REF) * (1+R1/R2) 0V I(L) Inductor Current 0A V(VCC) FAULT Hiccup Mode When the converter enters a fault mode, the SP6121 holds the high side and low side MOSFETs off for a finite period of time. Provided that the SP6121 is enabled, this time is set by the internal charge of the SS capacitor. In the event of an over-current condition, the current sense comparator sets the fault latch, which in turn discharge the internal SS capacitor, the COMP pin and holds the output drivers off. During this condition, the SP6121 stays off for the time it takes to discharge the COMP pin down below the 0.3V shutdown threshold. As soon as the COMP pin reaches 0.3V, the fault latch is reset and the SP6121 is allowed to attempt restart just like during a normal soft start cycle. The COMP pin has to charge back to 1V before any output switching can take place. At this point, the regulator attempts to restart normally by delivering short gate pulses to the output switches. If the over-current condition persists, the regulator will be kept off for the total time that it takes to charge the internal softstart capacitor to within 1V from the input supply voltage VCC plus the time required by the COMP voltage to cross the 1V threshold. This total time is typically several milli-seconds and minimizes thermal stress to the regulator components as the over-current condition persists. The waveforms that describe the hiccup mode operation are shown in Figure 2. V(ISET)-V(ISENSE) 160 mV 0V COMP Voltage 3V 0V V(VCC) SWN Voltage 0V TIME Figure 1. SP6121 Soft Start Waveforms Date: 5/25/04 1.0 V 0.3 V 0V V(VCC) PDRV Voltage 0V TIME Figure 2. SP6121 Hiccup mode waveforms (c) Copyright 2004 Sipex Corporation SP6121 Low Voltage, Synchronous Step Down PWM Controller 6 THEORY OF OPERATION: continued Over Current Protection Over current protection on the SP6121 is implemented through detection of an excess voltage condition across the high side PMOS switch during conduction. This is typically referred to as high side RDS(ON) detection and eliminates the need of an external sense resistor. The over current comparator charges an internal sampling capacitor each time V(ISET) - V(ISENSE) exceeds the 160mV (typ) internal threshold and the PDRV voltage is low. The discharge/charge current ratio on the sampling capacitor is about 2%. Therefore, provided that the over current condition persists, the capacitor voltage will be pumped up during each time PDRV switches low. This voltage will trigger an over current condition upon reaching a CMOS inverter threshold. There are many advantages to this approach. First, the filtering action of the gated S/ H scheme protects against false and undesirable triggering that could occur during a minor transient overload condition or supply line noise. Furthermore, the total amount of time to trigger the fault depends on the on-time of the PMOS switch. Ten, 1s pulses are equivalent to twenty, 500ns pulses or one, 10s pulse, however, depending on the period, each scenario takes a different amount of total time to trigger a fault. Therefore, the fault becomes an indicator of average power in the PMOS switch. Although the 160 mV internal threshold is fixed, the overall RDS(ON) detection voltage can be increased by placing a resistor from ISET to the source of the PMOS. A 30A sink current programs the additional voltage. In order for the current limit circuit to operate properly and accurately, the ISET and ISENSE pins must be Kelvin connected to the high side PMOS's source and drain pins. The 160mV threshold and 30A ISET current have 3300 ppm/C temperature coefficients in an effort to first order match the thermal characteristics of the RDS(ON) of the PMOS switch. It assumed that the SP6121 will be used in compact designs where there is a high amount of thermal coupling between the PMOS and the controller. Output Drivers The SP6121, unlike some other bipolar controller IC's, incorporates gate drivers with rail-torail swing that help prevent spurious turn on due to capacitive coupling. The driver stage consists of one high side PMOS, 4 driver, PDRV, and one low side, 4, NFET driver, NDRV, optimized for driving external power MOSFET's in a synchronous buck topology. The output drivers also provide gate drive non-overlap mechanism that provides a dead time between PDRV and NDRV transitions to avoid potential shootthrough problems in the external MOSFET's. Figure 3 shows typical waveforms for the output drivers. As with all synchronous designs, care must be taken to ensure that the MOSFETs are properly chosen for non-overlap time, enhancement gate drive voltage, "on" resistance RDS(ON), reverse transfer capacitance Crss, input voltage and maximum output current. GATE DRIVER TEST CONDITIONS 5V 90 % FALL TIME Vcc-2 V NON-OVERLAP PDRV(NDRV) 10 % 5V 90 % RISE TIME 2V 10 % NDRV(PDRV) V(VCC) PDRV Voltage 0V V(VCC) NDRV Voltage 0V V(VCC=VIN) SWN Voltage ~0V - V(Diode) V TIME Figure 3. SP6121 Output Driver Waveforms. Date: 5/25/04 SP6121 Low Voltage, Synchronous Step Down PWM Controller (c) Copyright 2004 Sipex Corporation 7 APPLICATION INFORMATION Inductor Selection There are many factors to consider in selecting the inductor including cost, efficiency, size and EMI. In a typical SP6121 circuit, the inductor is chosen primarily for value, saturation current and DC resistance. Increasing the inductor value will decrease output voltage ripple, but degrade transient response. Low inductor values provide the smallest size, but cause large ripple currents, poor efficiency and more output capacitance to smooth out the larger ripple current. The inductor must also be able to handle the peak current at the switching frequency without saturating, and the copper resistance in the winding should be kept as low as possible to minimize resistive power loss. A good compromise between size, loss and cost is to set the inductor ripple current to be within 20% to 40% of the maximum output current. The switching frequency and the inductor operating point determine the inductor value as follows: L= VOUT (V IN (max) - VOUT ) VIN (max) FS Kr I OUT ( max) duce considerable ac core loss, especially when the inductor value is relatively low and the ripple current is high. Ferrite materials, on the other hand, are more expensive and have an abrupt saturation characteristic with the inductance dropping sharply when the peak design current is exceeded. Nevertheless, they are preferred at high switching frequencies because they present very low core loss and the design only needs to prevent saturation. The power dissipated in the inductor is equal to the sum of the core and copper losses. To minimize copper losses, the winding resistance needs to be minimized, but this usually comes at the expense of a larger inductor. Core losses have a more significant contribution at low output current where the copper losses are at a minimum, and can typically be neglected at higher output currents where the copper losses dominate. Core loss information is usually available from the magnetic vendor. The copper loss in the inductor can be calculated using the following equation: 2 PL( Cu) = I L ( RMS ) RWINDING where; FS = switching frequency Kr = ratio of the peak to peak inductor ripple current to the maximum output current The peak to peak inductor ripple current is: where IL(RMS) is the RMS inductor current that can be calculated as follows: IL(RMS) = IOUT(max) 1+ 1 IPP 3 IOUT(max) ( ) 2 I PP = VOUT (VIN (max) - VOUT ) VI N (max) FS L Once the required inductor value is selected, the proper selection of core material is based on peak inductor current and efficiency requirements. The core material must be large enough not to saturate at the peak inductor current I PEAK = I OUT (max) + I PP 2 and provide low core loss at the high switching frequency. Low cost powdered iron cores have a gradual saturation characteristic but can introDate: 5/25/04 Output Capacitor Selection The required ESR (Equivalent Series Resistance) and capacitance drive the selection of the type and quantity of the output capacitors. The ESR must be small enough that both the resistive voltage deviation due to a step change in the load current and the output ripple voltage do not exceed the tolerance limits expected on the output voltage. During an output load transient, the output capacitor must supply all the additional current demanded by the load until the SP6121 adjusts the inductor current to the new value. Therefore the capacitance must be large enough so that the output voltage is held up SP6121 Low Voltage, Synchronous Step Down PWM Controller (c) Copyright 2004 Sipex Corporation 8 APPLICATIONS INFORMATION: Continued while the inductor current ramps up or down to the value corresponding to the new load current. Additionally, the ESR in the output capacitor causes a step in the output voltage equal to the ESR value multiplied by the change in load current. Because of the fast transient response provided by the SP6121 when exposed to output load transient, the output capacitor is typically chosen for ESR , not for capacitance value. The output capacitor's ESR, combined with the inductor ripple current, is typically the main contributor to output voltage ripple. The maximum allowable ESR required to maintain a specified output voltage ripple can be calculated by: Panasonic offers the SP series of specialty polymer aluminum electrolytic surface mount capacitors. These capacitors have a lower ESR than tantalum capacitors, reducing the total number of capacitance required for a given transient response. Input Capacitor Selection The input capacitor should be selected for ripple current rating, capacitance and voltage rating. The input capacitor must meet the ripple current requirement imposed by the switching current. In continuous conduction mode, the source current of the high-side MOSFET is approximately a square wave of duty cycle VOUT/ VIN. Most of this current is supplied by the input bypass capacitors. The RMS value of input capacitor current is determined at the maximum output current and under the assumption that the peak to peak inductor ripple current is low, it is given by: ICIN(rms) = IOUT(max) D(1 - D) The worse case occurs when the duty cycle D is 50% and gives an RMS current value equal to IOUT/2. Select input capacitors with adequate ripple current rating to ensure reliable operation. The power dissipated in the input capacitor is: 2 P = ICIN ( rms ) R ESR( CIN ) CIN RESR VOUT I PP where; VOUT = peak to peak output voltage ripple IPP = peak to peak inductor ripple current The total output ripple is a combination of the ESR and the output capacitance value and can be calculated as follows: VOUT = IPP (1 - D) COUTFS ( ) 2 + (IPPRESR)2 where; D = duty cycle equal to VOUT/VIN COUT = output capacitance value Recommended capacitors that can be used effectively in SP6121 applications are: low-ESR aluminum electrolytic capacitors, OS-CON capacitors that provide a very high performance/ size ratio for electrolytic capacitors and lowESR tantalum capacitors. AVX TPS series and Kemet T510 surface mount capacitors are popular tantalum capacitors that work well in SP6121 applications. POSCAP from Sanyo is a solid electrolytic chip capacitor that has low ESR and high capacitance. For the same ESR value, POSCAP has lower profile compared with tantalum capacitor. This can become a significant part of power losses in a converter and reduce the overall energy transfer efficiency. The input voltage ripple primarily depends on the input capacitor ESR and capacitance. Ignoring the inductor ripple current, the input voltage ripple can be determined by: VIN = IOUT(MAX)RESR(CIN) + IOUT(MAX)VOUT(VIN - VOUT) FSCINVIN2 The capacitor type suitable for the output capacitors can also be used for the input capacitors. However, exercise extra caution when tantalum capacitors are considered. Tantalum capacitors are known for catastrophic failure when exposed to surge current, and input capacitors are prone to such surge current when power Date: 5/25/04 SP6121 Low Voltage, Synchronous Step Down PWM Controller (c) Copyright 2004 Sipex Corporation 9 supplies are connected `live' to low impedance power sources. Certain tantalum capacitors, such as AVX TPS series, are surge tested. For generic tantalum capacitors, use 2:1 voltage derating to protect the input capacitors from surge fall-out. MOSFET Selection The SP6121 drives a PMOS MOSFET on the high side and an NMOS MOSFET synchronous rectifier on the low side. Using a PMOS switch on the high side negates the need for an external charge pump and simplifies the application circuit. The losses associated with MOSFETs can be divided into conduction and switching losses. Conduction losses are related to the on resistance of MOSFETs, and increase with the load current. Switching losses occur on each on/off transition when the MOSFETs experience both high current and voltage. Since the bottom MOSFET switches current from/to a paralleled diode (either its own body diode or a Schottky diode), the voltage across the MOSFET is no more than 1V during switching transition. As a result, its switching losses are negligible. The switching losses are difficult to quantify due to all the variables affecting turn on/off time. However, making the assumption that the turn on and turn off transition times are equal, the transition time can be approximated by: tT = CISSVIN , IG where; CISS is the PMOS's input capacitance, or the sum of the gate-to-source capacitance, CGS, and the drain-to-gate capacitance, CGD. This parameter can be directly obtained from the MOSFET's data sheet IG is the gate drive current provided by the SP6121 (approximately 1A at VIN=5V) and VIN is the input supply voltage. Therefore an approximate expression for the switching losses associated with the high side MOSFET can be given as: PSH(max) = (VIN(max) + VF)IOUT(max)tTFS , where; tT = the switching transition time VF = free wheeling diode drop Date: 5/25/04 APPLICATIONS INFORMATION: Continued Switching losses need to be taken into account for high switching frequency, since they are directly proportional to switching frequency. The conduction losses associated with top and bottom MOSFETs are determined by PCH(max) = RDS(ON)IOUT(max) D PCL(max) = RDS(ON)IOUT(max) (1 - D), where; PCH(max) = conduction losses of the high side MOSFET PCL(max) = conduction losses of the low side MOSFET RDS(ON) = drain to source on resistance. The total power losses of the top MOSFET are the sum of switching and conduction losses. For synchronous buck converters of efficiency over 90%, allow no more than 4% power losses for high or low side MOSFETs. For input voltages of 3.3V and 5V, conduction losses often dominate switching losses. Therefore, lowering the RDS(ON) of the MOSFETs always improves efficiency even though it gives rise to higher switching losses due to increased CISS . Total gate charge is the charge required to turn the MOSFETs on and off under the specified operating conditions (VGS and VDS). The gate charge is provided by the SP6121 gate drive circuitry. (At 500kHz switching frequency, the gate charge is the dominant source of power dissipation in the SP6121.) At low output levels, this power dissipation is noticeable as a reduction in efficiency. The average current required to drive the high side and low side MOSFETs is: IG(av) = QGHFS + QGLFS, where; QGH = gate charge of PMOS QGL = gate charge of NMOS Considering that the gate charge current comes from the input supply voltage VIN, the power dissipated in the SP6121 due to the gate drive is: PGATE DRIVE = VINIG(av) RDS(ON) varies greatly with the gate driver voltage. The MOSFET vendors often specify RDS(ON) on multiple gate to source voltages (VGS), as (c) Copyright 2004 Sipex Corporation 2 2 SP6121 Low Voltage, Synchronous Step Down PWM Controller 10 APPLICATIONS INFORMATION: Continued well as provide typical curve of RDS(ON) versus VGS. For 5V input, use the RDS(ON) specified at 4.5V VGS. At the time of this publication, vendors, such as Fairchild, Siliconix and International Rectifier, have started to specify RDS(ON) at VGS less than 3V. This has provided necessary data for designs in which these MOSFETs are driven with 3.3V and made it possible to use SP6121 in 3.3V only applications. Thermal calculation must be conducted to ensure the MOSFET can handle the maximum load current. The junction temperature of the MOSFET, determined as follows, must stay below the maximum rating. Schottky diode alleviates this noise and additionally improves efficiency thanks to its low forward voltage. The reverse voltage across the diode is equal to input voltage, and the diode must be able to handle the peak current equal to the maximum load current. The power dissipation of the Schottky diode is determined by PDIODE = 2VFIOUTTNOLFS where; TNOL = non-overlap time between PDRV and NDRV. VF = forward voltage of the Schottky diode. TJ ( max) = T A (max) + PMOSFET (max) R JA , COMP (R) where; TA(max) = maximum ambient temperature PMOSFET(max) = maximum power dissipation of the MOSFET RJA = junction to ambient thermal resistance. RJA of the device depends greatly on the board layout, as well as device package. Significant thermal improvement can be achieved in the maximum power dissipation through the proper design of copper mounting pads on the circuit board. For example, in a SO-8 package, placing two 0.04 square inches copper pad directly under the package, without occupying additional board space, can increase the maximum power from approximately 1 to 1.2W. For DPAK package, enlarging the tap mounting pad to 1 square inches reduces the RJA from 96C/W to 40C/W. Schottky Diode Selection When paralleled with the bottom MOSFET, an optional Schottky diode can improve efficiency and reduce noise. Without this Schottky diode, the body diode of the bottom MOSFET conducts the current during the non-overlap time when both MOSFETs are turned off. Unfortunately, the body diode has high forward voltage and reverse recovery problem. The reverse recovery of the body diode causes additional switching noises when the diode turns off. The Date: 5/25/04 R1 C2 C1 SP6121 Figure 4. The RC network connected to the COMP pin provides a pole and a zero to control loop. Loop Compensation Design The goal of loop compensation is to manipulate loop frequency response such that its gain crosses over 0db at a slope of -20db/dec. The SP6121 has a trans-conductance error amplifier and requires the compensation network to be connected between the COMP pin and ground, as shown in Figure 4. The first step of compensation design is to pick the loop crossover frequency. High crossover frequency is desirable for fast transient response, but often jeopardize the system stability. Crossover frequency should be higher than the ESR zero but less than 1/5 of the switching frequency. The ESR zero is contributed by the ESR SP6121 Low Voltage, Synchronous Step Down PWM Controller (c) Copyright 2004 Sipex Corporation 11 APPLICATIONS INFORMATION: Continued associated with the output capacitors and can be determined by fZ(ESR) = 1 2COUTRESR Gain -20db/dec To fine-tune the compensation, it is necessary to physically measure the frequency response using a network analyzer. Crossover frequency of 20kHz is a sound first try if low ESR tantalum capacitors or poscaps are used at the output. The next step is to calculate the complex conjugate poles contributed by the LC output filter, 1 fP(LC) = 2 LCOUT The open loop gain of the whole system can be divided into the gain of the error amplifier, PWM modulator, buck converter, and feedback resistor divider. In order to crossover at the selected frequency fco, the gain of the error amplifier has to compensate for the attenuation caused by the rest of the loop at this frequency. In the RC network shown in Figure 4, the product of R1 and the error amplifier transconductance determines this gain. Therefore, R1 can be determined from the following equation that takes into account the typical error amplifier transconductance, reference voltage and PWM ramp built into the SP6121. 975VOUT fCO fZ(ESR) R1 = VIN fP(LC)2 In Figure 4, R1 and C1 provides a zero fZ1 which needs to be placed at or below fP(LC). If fZ1 is made equal to fP(LC) for convenience, the value of C1 can be calculated as C1 = 1 2fP(LC)R1 -40db/dec Loop -20db/dec f -20db/dec Error Amplifier -20db/dec f Figure 5. Frequency response of a stable system and its error amplifier. The optional C2 generates a pole fP1 with R1 to cut down high frequency noise for reliable operation. This pole should be placed one decade higher than the crossover frequency to avoid erosion of phase margin. Therefore, the value of the C2 can be derived from 1 C2 = 20fCOR1 Figure 5 illustrates the overall loop frequency response and frequency of each pole and zero. Date: 5/25/04 Overcurrent Protection Over current protection on the SP6121 is implemented through detection of an excess voltage condition across the high side PMOS switch during conduction. This is typically referred to as high side RDS(ON) detection. By using the RDS(ON) of Q1 to measure the output current, the current limit circuit eliminates the sense resistor that would otherwise be required and the corresponding loss associated with it. This improves the overall efficiency and reduces the number of components in the power path benefiting size and cost. RDS(ON) sensing is by default inaccurate and is primarily meant to protect the power supply during a fault condition. The overcurrent trip point will vary from unit to unit as the RDS(ON) of Q1 varies. The SP6121 provides a built-in 160mV threshold between the ISET and ISENSE pins. If a current limit threshold in excess of 160mV is required, an external programming resistor, RSET can be added between ISET pin and VIN as shown in Figure 6. (c) Copyright 2004 Sipex Corporation SP6121 Low Voltage, Synchronous Step Down PWM Controller 12 APPLICATIONS INFORMATION: Continued both the 30A sink current present at the ISET pin V and the 160mV built-in current limit threshold SP6121 RSET C have been designed with a positive temperature 160mV I -+ + coefficient of about 0.33%/C to provide first order Q1 PDRV correction for current limit versus temperature. I 30A L1 V This compensation relies on the high amount of C thermal coupling that typically exists between the high side switch Q1 and the SP6121 due to the Q2 NDRV compact size of the power supply. With this first order compensation, the current limit trip point does not need to be set to an increased level at room Figure 6. Current Limit Setting temperature to guarantee a desired output current The value of RSET can be properly chosen based on level at higher temperatures. the desired current limit point IMAX and the internal 30A pull down current available on the ISET pin Output Voltage Program according to the following expression: As shown in Figure 7(A), the voltage divider connecting to the VFB pin programs the output IMAXRDS(ON) - 160mV RSET = voltage according to ISET IN IN SET SENSE OUT OUT where, ISET = 30A (typ) sink current from the ISET pin. Kelvin-Sense connections should be made directly at the drain and source of Q1. The RDS(ON) sensing scheme implemented in the SP6121 provides two additional features that enhance the performance of the overcurrent function. First, an internal sample and hold filter connected after the main current-sense comparator, prevents that noise spikes or very short and mild overload conditions, that could occur during a load transient, spuriously activate the current limit circuitry. This typically eliminates the need of using any external filtering that would be otherwise required. Additionally, since the RDS(ON) has a positive temperature coefficient, VOUT > 1.25V VOUT = 1.25 1 + R1 R2 where 1.25V is the internal reference voltage. Select R2 in the range of 10k to 100k, and R1 can be calculated using R2(VOUT - 1.25) R1 = 1.25 For output voltage less than 1.25V, a simple circuit shown in Figure 7(B) can be used in which VREF is an external voltage reference greater than 1.25V. For simplicity, use the same resistor value for R1 and R2, then R3 is determined as follows, (VREF - 1.25)R1 R3 = 2.5V - VOUT VREF VOUT < 1.25V ( ) (R) R1 VFB (R) R3 VFB R1 SP6121 SP6121 R2 R2 A B Figure 7(A) A voltage divider connected to the VFB pin programs the output voltage. (B) A simple circuit using one external voltage reference programs the output voltages less than 1.25V. Date: 5/25/04 SP6121 Low Voltage, Synchronous Step Down PWM Controller (c) Copyright 2004 Sipex Corporation 13 LAYOUT GUIDELINE PCB layout plays a critical role in proper function of the converters and EMI control. In switch mode power supplies, loops carrying high di/dt give rise to EMI and ground bounces. The goal of layout optimization is to identify these loops and minimize them. It is also crucial on how to connect the controller ground such that its operation is not affected by noise. The following guideline should be followed to ensure proper operation. 1. A ground plane is recommended for minimizing noises, copper losses and maximizing heat dissipation. 2. Begin the layout by placing the power components first. Orient the power circuitry to achieve a clean power flow path. If possible make all the connections on one side of the PCB with wide, copper filled areas. 3. Connect the ground of feedback divider and compensation components directly to the GND pin of the IC using a dedicated ground trace. Then connect this pin as close as possible to the ground of the output capacitor. 4. The VCC bypass capacitor should be right next to the VCC and GND pins. 5. The trace connecting the feedback resistors to the output should be short, direct and far away from the switch node, and switching components. 6. Minimize the trace between PDRV/NDRV and the gates of the MOSFETs to reduce the impedance driving the MOSFETs. This is especially important for the bottom MOSFET that tends to turn on through its Miller capacitor when the switch node swings high. 7. Minimize the loop composed of input capacitors, top/bottom MOSFETs and Schottky diode. This loop carries high di/dt current. Also increase the trace width to reduce copper losses. 8. Maximize the trace width of the loop connecting the inductor, output capacitors, Schottky diode and bottom MOSFET. 9. ISET and ISENSE connections to Q1 for current limiting must be make using Kelvin connections. Date: 5/25/04 SP6121 Low Voltage, Synchronous Step Down PWM Controller (c) Copyright 2004 Sipex Corporation 14 PACKAGE: 8 PIN NSOIC D e E/2 B E1/2 E1 B E SEE VIEW C 1 INDEX AREA (D/2 X E1/2) b O1 TOP VIEW L2 Seating Plane O1 L L1 O VIEW C Gauge Plane A A2 Seating Plane A1 SIDE VIEW 8 Pin NSOIC (JEDEC MS-012, AA - VARIATION) DIMENSIONS in (mm) b c BASE METAL SECTION B-B WITH PLATING SYMBOL A A1 A2 b c D E E1 e L L1 L2 O O1 MIN NOM MAX 1.75 1.35 0.25 0.10 1.25 1.65 0.31 0.51 0.17 0.25 4.90 BSC 6.00 BSC 3.90 BSC 1.27 BSC 0.40 1.27 1.04 REF 0.25 BSC 8 0 5 15 8 PIN NSOIC Date: 5/25/04 SP6121 Low Voltage, Synchronous Step Down PWM Controller (c) Copyright 2004 Sipex Corporation 15 ORDERING INFORMATION Part Number Operating Temperature Range Package Type SP6121CN ............................................... 0C to +70C ........................................... 8-Pin nSOIC SP6121CN/TR ......................................... 0C to +70C .......................................... 8-Pin nSOIC Available in lead free packaging. To order add "-L" suffix to part number. Example: SP6121CN/TR = standard; SP6121CN-L/TR = lead free /TR = Tape and Reel Pack quantity is 2500 for NSOIC. Corporation ANALOG EXCELLENCE Sipex Corporation Headquarters and Sales Office 233 South Hillview Drive Milpitas, CA 95035 TEL: (408) 934-7500 FAX: (408) 935-7600 Sipex Corporation reserves the right to make changes to any products described herein. Sipex does not assume any liability arising out of the application or use of any product or circuit described herein; neither does it convey any license under its patent rights nor the rights of others. Date: 5/25/04 SP6121 Low Voltage, Synchronous Step Down PWM Controller (c) Copyright 2004 Sipex Corporation 16 |
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